Flexible low current oscillator for multiphase operations

ABSTRACT

A method for generating an oscillator signal uses a multiphase oscillator having a plurality of input stages and a reference stage. Each input stage produces an input stage voltage that represents a phase for the oscillator. The input stage voltages produced by each of the input stages are compared to a reference voltage produced by the reference stage. An input stage having a maximum input stage voltage is selected and an output of the selected input stage having the maximum input stage voltage is changed. A current need of the oscillator is detected with a negative feedback loop coupled to the reference stage. An appropriate supply current is provided to each input stage with the negative feedback loop.

CLAIM OF PRIORITY

This application is a continuation of and claims the priority benefit of commonly-assigned, co-pending U.S. patent application Ser. No. 12/488,413 entitled “FLEXIBLE LOW CURRENT OSCILLATOR FOR MULTIPHASE OPERATIONS” to Behzad Mohtashemi, filed Jun. 19, 2009, the entire disclosures of which are incorporated herein by reference.

FIELD OF THE INVENTION

This invention generally relates to an electrical circuit for generating a cyclic signal and more particularly to a flexible oscillator with low transistor count, ability to oscillate to very high frequency, and low current consumption, which can be used in multiphase operations.

BACKGROUND OF THE INVENTION

Within nearly every electronic subsystem is some form of waveform generator that produces cyclical waveforms. The waveform generator is oftentimes referred to as an oscillator. Depending on the application, an oscillator can be used to source regularly spaced pulses or clock signals. Oscillators are oftentimes rated depending on their stability and accuracy, frequency adjustability (i.e., tunability), gain of active circuit, start-up time, power consumption, etc.

A type of oscillator commonly known as a relaxation oscillator is the most commonly used architecture for lower frequency oscillator designs. FIG. 1A is an electrical schematic diagram illustrating an example of a conventional relaxation oscillator. As shown in FIG. 1A, a relaxation oscillator 100 includes a capacitor C₁₀, a switching device SW₁₀ such as a field effect transistor, a comparator 102, a current source I₁₀ and a one-shot timer 104.

A voltage reference V_(Ref) is connected to the − input of the comparator 102. A first terminal of the capacitor C₁₀ is connected to the + input of the comparator 102. The second terminal of the capacitor C₁₀ is grounded. The output of the comparator 102 is electrically connected to the input of the one-shot timer 104, the output of which is electrically coupled to a control terminal of the switch device SW₁₀. The switch device SW₁₀ is electrically connected between the first terminal of the capacitor C₁₀ and ground and is used for discharging the frequency-determining capacitor C₁₀. As shown in the voltage versus time graph of FIG. 1B, the voltage on the capacitor is more or less saw-toothed in shape with a short flat spot 101 between successive saw-teeth.

For these particular oscillators, the frequency is limited by the speed of the comparator 102. As current from the source I₁₀ charges the capacitor C₁₀ the voltage at the (+) input of the comparator 102 eventually reaches the reference voltage V_(Ref) and turns on comparator 102. That triggers the one shot 104 to open the switch SW₁₀, which discharges the capacitor C₁₀ and sets the voltage back to zero. The one shot 104 keeps the switch SW₁₀ on long enough to completely discharge the capacitor C₁₀ so that the output is not erratic. In order to ensure that the capacitor C₁₀ completely discharges, there will be a delay time 101 between successive saw-teeth.

For low frequency, delay times in the comparator 102 are relatively small. But if switching is to be done at high frequency, the delay time becomes large with respect to each saw-tooth cycle. Also, at high frequency, e.g. 5 MHz, a high switching current, e.g., 1 mA is needed. The current source that charges up the capacitor C₁₀ coupled to the one shot 104 increases as the input voltage increases. So, for example, at 5.0V the current source I₁₀ may be able to deliver 5 μA, and at 2.5 it can deliver only 1 μA. Thus, at 5.0V, for example, the capacitor charges up 5 times faster than at 2.5V. So, at 5.0V, the flat response is much less than what it is at 2.5V. In a low frequency where switching frequency is at 500 KHz, it might not be a problem for the one shot 104 to vary from 50 ns to 70 ns, but in a high frequency oscillation, e.g., greater than 3 MHz, it matters very much to have variations from 50 ns, to 100 ns.

Since one shot properties vary with supply voltage, the flat spot between sawtooth waves changes with supply voltage and also varies with temperature, which is not desirable. As a result, relaxation oscillators are only good for frequencies of about 1 MHz and below. In addition, these oscillators are not well suited for multiphase systems. In a conventional multiphase system, each phase need its own particular oscillator and comparator. Furthermore, the conventional comparator is not cheap and the oscillator consumes a lot of current. High current consumption is undesirable in many applications, such as portable devices.

It is within this context that embodiments of the present invention arise.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a circuit diagram of a conventional relaxation oscillator.

FIG. 1B shows a signal profile of the circuit of FIG. 1A.

FIGS. 2A-2B are circuit diagrams illustrating the principles of operation of the oscillators and the reference stage according to embodiments of the present invention.

FIG. 3A is a circuit diagram of an NMOS two phase oscillator according to an embodiment of present invention.

FIG. 3B is a circuit diagram of a PMOS two phase oscillator according to an embodiment of present invention.

FIGS. 3C-3D show signal profiles at nodes N₁ and N₂ in the electrical circuits of the type shown in FIGS. 3A-3B respectively.

FIG. 3E shows the combined signal profiles of nodes N₁ and N₂.

FIG. 3F shows a circuit diagram of a NMOS two phase oscillator with an alternative reference stage.

FIG. 4A is a circuit diagram of a three phase oscillator according to an embodiment of the present invention.

FIGS. 4B-4D shows signal profile at nodes N₁, N₂ and N₃ of FIG. 4A at an oscillation frequency of 50 MHz.

FIG. 4E shows signal profile at nodes N₁, N₂ and N₃ of FIG. 4A at an oscillation frequency higher than 50 MHz.

FIG. 5A is a circuit diagram of a four phase oscillator according to an embodiment of the present invention.

FIGS. 5B-5E shows signal profile at nodes N₁, N₂, N₃ and N₄ of FIG. 5A at an oscillation frequency higher than 50 MHz.

FIG. 5F shows signal profile at nodes N₁, N₂, N₃ and N₄ of FIG. 5A at an oscillation frequency higher than 50 MHz.

DESCRIPTION OF THE SPECIFIC EMBODIMENTS

Embodiments of the present invention are directed to oscillators that overcome the disadvantages associated with prior art relaxation oscillators. Oscillator designs in accordance with embodiments of the present invention eliminate the need for a conventional comparator. The oscillator is generally comprised of one or more differential input stages that may have as many input stages as the number of phases needed with a small number of active components. Each differential stage acts as a maximum selector stage. As used herein, the term “maximum selector” refers is a circuit having multiple input stages (sometimes referred to herein as “phase stages”) that chooses the maximum voltage of the voltages of the different input stages. Selection of a given input stage by the maximum selector causes an output of the selected input stage to change. The current supplied to a given input stage may be varied as the oscillator's needs change. For example, a negative feedback loop coupled to the reference stage may be configured to detect a current need of the oscillator and provide an appropriate supply current to each input stage to meet the detected current need.

According to an embodiment of the invention, an oscillator apparatus may comprise a reference stage and two or more phase stages including a first stage and a last stage. The reference stage has a reference current source and a first reference transistor having a gate coupled to a voltage reference and a drain coupled to the reference current source. By way of example, and not by way of limitation, each of the phase stages may include a transistor, first and second current sources, a capacitor, a switch, and a logic block. The transistor has a drain coupled to the first current source, a gate coupled to a node and a source coupled to a source of the reference transistor in the reference stage. The capacitor has a first terminal coupled to the node and a second terminal connected to ground. The switch is coupled between the first and second terminals of the capacitor. The second current source is coupled to the node.

Basically, the logic block for a given phase stage is used to latch on the switch for that phase stage, and reset the switch for the next phase stage. In other words, when a phase stage is selected by the maximum selector, the logic block deactivates the selected phase stage, and activates the next phase stage (which corresponds to the next phase of the oscillator). The logic block for each phase stage may be formed in many different ways. An example of a logic block is shown which includes an inverter and a set-reset latch. An input of the inverter for a given phase stage is coupled to the drain of the transistor for that phase stage. The output of the inverter for a given phase stage is coupled to a set input for the latch for that phase stage. An output of the latch for a given phase stage is coupled to the switch for that phase stage. The output of the inverter for a given phase stage is also coupled to the reset input of the set-reset latch in a subsequent phase stage. The output of the inverter for the last stage is coupled to the reset input of the set-reset latch in the first stage.

FIG. 2A is a circuit diagram illustrating an input stage (or phase stage) of a phase oscillator 200. As shown in FIG. 2A, the oscillator 200 includes a reference stage 202 and a phase stage 204. The reference stage 202 includes a reference field effect transistor T_(ref) having a drain D_(ref) electrically coupled to a first reference current source I_(ref), gate G_(ref) electrically connected to a voltage reference V_(ref) and a source S_(ref) coupled to an input of a second reference current source I_(ref)′, which may supply a reference current of up to, e.g., about twice the current supplied by the first reference current source I_(ref). By way of example, the reference transistor T_(ref) may be a metal oxide semiconductor (MOS) device. By way of example, the first reference current source I_(ref) may supply a current of about 1 μA and the second reference current source I_(ref)′ may supply a current of about 2 μA and the reference voltage V_(ref) may be about 1.2 volts.

The phase stage 204 also includes a transistor T₁ having a source S₁, a gate G₁, and a drain D₁. The source S₁ is connected to the source S_(ref) of the reference transistor T_(ref), at a first junction J₁. The gate G₁ is coupled to a second junction J₂ and the drain D₁ is connected to the output of a current source I₁ that supplies a current roughly the same as the current supplied by the first reference current source I_(ref). The oscillator 200 also includes a capacitor C having a first terminal connected to the junction J₂ and a second terminal that is grounded. The oscillator 200 also includes a switch SW that permits current to flow between the second junction J₂ and ground to discharge capacitor C when a control electrode of the switch SW receives an input trigger signal. Input sides of current sources I_(ref), I₁ and a third current source I₁′ are coupled to a third junction J₃. The output of the third current source I₁′ is coupled to the junction J₂.

Operation of the oscillator 200 may proceed as follows. Suppose initially, the trigger signal turns switch SW on, therefore the transistor T₁ is off since the gate G₁ is coupled to ground. The transistor T_(ref) is on due to the reference voltage V_(ref) applied to the gate G_(ref) of the reference transistor T_(ref). The larger current from the second reference source I_(ref)′ pulls the drain D_(ref) of the reference transistor T_(ref) down, but the smaller current from the source I₁ pulls up the drain D₁ of transistor T₁.

After the switch SW is turned off, the current from the third current source I₁′ charges up the capacitor C and slowly turns the transistor T₁ on and the reference transistor T_(ref) off as the gate G₁ of the transistor T₁ goes above the reference voltage V_(ref). The current source L_(ref)′ pulls down the voltage at the drain D₁ of the transistor T₁ and pulls up the voltage at the first junction J₁ and at the drain D_(ref) of the reference transistor T_(ref). This situation remains until the switch SW turns on again. Digital output signals may be obtained from the voltages at the drains D₁ and D_(ref) of the transistors T₁ and T_(ref). In order for the oscillator to operate at a relatively high rate, the current from the second reference current source I_(ref)′ should be relatively large, as discussed above.

Alternatively, instead of using a constant current I_(ref)′, a current control may use negative feedback based on the voltage between the first reference source I_(ref) and the reference transistor T_(ref) (which may be referred to as the first reference transistor) may be used to control the current from the transistor T₁ through the junction J₁. For example, FIG. 2B is an electrical diagram of an alternative oscillator 210, that is a variation on the oscillator 200 shown in FIG. 2A. In addition to the components described above with respect to FIG. 2A, a reference stage 212 of the oscillator 210 includes a current control in the form of second reference transistor T_(ref)′ (e.g., a MOS transistor) having a drain D_(ref)′ coupled to first junction J₁, a gate G_(ref)′ coupled to the drain D_(ref) of first reference transistor T_(ref) and a source S_(ref)′ connected to ground. It is noted that in this example illustrated in FIG. 2B, the reference transistor T_(ref)′ is an N-type (or n-channel) device. Alternatively, the reference transistor T_(ref)′ may be a P-type (or p-channel) device, in which case the source S_(ref)′ may be coupled to a common voltage V_(cc). The first reference current source I_(ref) acts as a fixed reference current source. If the voltage at junction J₁ goes high (e.g., when T₁ turns on) the drain D_(ref) of first reference transistor T_(ref) and gate G_(ref)′ of the second reference transistor T_(ref)′ to be pulled high, which turns the second reference transistor T_(ref)′ more on which increases the supply current I_(s) and pulls junction J₁ back down. Without the negative feedback, an increase of voltage at junction J₁ would cause the first reference transistor T_(ref) to turn off because its gate-to-source voltage (i.e., V_(ref) minus the voltage at junction J₁) decreases. Conversely, if the voltage at junction J₁ goes low it pulls the gate G_(ref)′ of the transistor T_(ref)′ low, which pulls the junction J₁ back up. If the second reference transistor T_(ref)′ cannot supply sufficient current, it pulls up the voltage at the drain D_(ref) of the transistor T_(ref) and at the junction J₁ which causes the second reference transistor T_(ref)′ to open more until it provides the appropriate supply current I_(s). This way, the first reference transistor T_(ref) is always on and allows the current supplied by supply I_(ref) to go through, no matter what the input is from transistor T₁. The second reference transistor T_(ref)′ changes the amount of current it allows through such that the value of current source I_(ref) always flows through reference transistor T_(ref). As a result of I_(ref), T_(ref), and T_(ref′), there is a negative feedback in the circuit of FIG. 2B. Because of this negative feedback, the supply current I_(s) of the circuit 210 changes to meet the current demand of the system.

The secondary reference transistor T_(ref)′ acts as a variable resistor that throttles the current of the reference current source I_(ref) in a negative feedback manner in response to changes in the current through junction J₁. Alternatively, another device capable of delivering a negative feedback loop such as an operational amplifier or variable resistor may be used as the current control to adjust the supply current I_(s) in response to changes in the voltage at junction J₁ to provide the desired negative feedback control of the current through the junction J₁.

Two or more phase stages of the type shown in FIG. 2A, 2B may be combined together with a reference block of the type shown in FIG. 2B to provide a multiple phase oscillator according to an embodiment of the present invention. By way of example, and not by way of limitation, FIG. 3A is an electrical diagram illustrating a two phase oscillator 300 according to an embodiment of the present invention. As shown in FIG. 3A, the oscillator 300 includes two phase stages 302, 304 and one reference stage 306. The reference stage 306 is similar to the reference stage 212 of the oscillator 210 described above in FIG. 2B.

By way of example, the first phase stage 302 may include a n-type MOS transistor T₁ having a drain D₁ coupled to a current source I₁, which may supply a current of about 1 uA through a junction J₁, a gate G₁ coupled to a first node N₁, and a source S₁ coupled to a junction J₇ of the reference stage 306. The first phase stage 302 also includes a capacitor C₁ having a first terminal coupled to secondary current source I₁′ through the first node N₁ and a second terminal is connected to the ground. A switch SW₁, which can be an n-type MOS transistor, is connected in parallel to the capacitor C₁ with a source grounded and having a gate coupled to an output of a latch L₁. The first phase stage 302 may further include a logic block 303 which includes an inverter INV₁ having an input coupled to the transistor T₁ at the junction J₁ and an output coupled to an input of the latch L₁. A one-shot timer OS₁ may be optionally coupled between the inverter INV₁ and the latch L₁. In such a case, the output of the inverter INV₁ may be regarded as being indirectly coupled to the latch L₁. The logic block shown here is only an example and may be replaced with a similarly functioning circuit block.

The second phase stage 304 includes similar components as the first phase stage 302. The second phase stage 304 may include a n-channel MOS transistor T₂ having a drain D₂ coupled to a current source I₂ of about 1 uA through a junction J₂, a gate G₂ coupled to secondary current source I₂′ through a second node N₂, and a source S₂ coupled to the junction J₇. The second phase stage 304 also includes a capacitor C₂ that is of equal capacitance to the capacitor C₁. The capacitor C₂ has a first terminal coupled to the second node N₂ and a second terminal is connected to the ground. A switch SW₂, which can be a n-channel MOS transistor, is coupled to the capacitor C₂, and includes a source connected to the ground and a gate coupled to an output of a latch L₂. The second phase stage 304 may further include a logic block 305 which includes an inverter INV₂ having an input coupled to the transistor T₂ at the junction J₂ and an output coupled to an input of the latch L₂. A one-shot timer OS₂ is optionally coupled between the inverter INV₂ and the latch L₂. The one-shot timers OS₁, OS₂ provide digital output signals if needed.

The reference stage 306 may operate in a manner similar to that of reference stage 212 as described above with respect to FIG. 2B. The reference stage 306 may operate with or without feedback control of the current. In the example shown in FIG. 3A, the second reference transistor T_(ref)′ provides negative feedback control of the current through the junction J₇ from the transistors T₁, T₂, and T_(ref).

As shown in FIG. 3A, the output of the one-shot timer OS₁ is connected to the set input S of the latch L₁ and to the Reset input R of the latch L₂. Similarly, the output of one-shot timer OS₂ is connected to a Set input S of the latch L₂ and a Reset input R of the latch L_(f). It is noted that if the one-shot timers are not used, the outputs of the inverters INV₁, INV₂ would be coupled directly to the latches.

The circuit 300 operates as a maximum selector that chooses as its output the maximum of the voltages of the phase stages 302, 304. In the example depicted in FIG. 3A, there are three voltages in the system, V_(ref), and the voltages at nodes N₁, N₂ (referred to as V_(n1), and V_(n2), respectively). The architecture shown in FIG. 3A always picks the largest of the voltages V_(n1), and V_(n2) and V_(ref). For example, if initially V_(ref) is 1.2V, and both N₁ and N₂ are at 0 volts. As soon as either V_(n1) or V_(n2) has risen above V_(ref), it is recognized by the architecture shown and it will be selected. For example, if V_(n2) has risen above V_(ref), then phase stage 304 is selected, and the transistor T₂ turns on. Because of the negative feedback in the reference stage 306, the reference transistor T_(ref) still stays on as will be explained below. With the transistor T₂ on, the voltage at J₁ is pulled low, which triggers the logic unit 305.

The supply current I_(s) is not fixed and it varies with the need of the system. In FIG. 3A, the arrangement of T_(ref) and T_(ref)′ provides a negative feedback, which means it will supply the required current for the intended architecture. The negative feedback ensures that T_(ref) always allows the value of the reference current source I_(ref) through, but also adjusts to provide the demand for current from the rest of the system. For example, when phase stage 304 is selected, the second reference transistor T_(ref)′ allows the supply current I_(s) through in the amount of I_(ref) plus I₂. However, when either of the transistors shown, T₁ or T_(z), turns on, there are parasitic capacitances that must be charged before these transistors can turn on. So the more current available, the faster these capacitances can be charged up and the quicker the transistors can turn on. During that transient, the supply current I_(s) will be increased due to the negative feedback on T_(ref)′. The negative feedback detects the current demand of the system, and provides the supply current I_(s) accordingly. Thus the negative feedback enables a more efficient, robust, and faster oscillator circuit. No matter how many stages are used, and what the transient current demand is, the negative feedback ensures that the appropriate supply current I_(s) is provided.

FIG. 3B is an electrical diagram showing an alternative two phase oscillator 310, which includes similar components as the oscillator 300, except the transistors T₁, T_(z), T_(ref) and T_(ref)′ and the switches SW₁ and SW₂ are p-channel MOS transistors. In this embodiment, the second terminals of the capacitors C₁, C₂, the sources of the switches SW₁, SW₂ and the source of the secondary reference transistor T_(ref)′ are coupled to a voltage Vcc instead of grounded. In other words, the positions of the Vcc and the ground are switched and so are the directions of the current sources I_(f), I₁′, I₂, I₂′ and I_(ref) in circuit 310, as compared to circuit 300 of FIG. 3A. In the logic portion of each phase stage for two phase oscillator 310, the inverter is placed between the latch and the switch, rather than between the one shot and the junction (e.g., J₁).

Speed of oscillation of the oscillators 300 and 310 is regulated by the amount of current provided by current sources I₁′, I₂′, and size of the capacitors C₁, C₂. The oscillators 300 and 310 can be used in dual phase converters.

The oscillators 300, 310 may operate as follows. At an initial state the switch SW₁ is off and the switch SW₂ is latched on by latch L₂. As a result, the transistor T₂ is off. Initially, the voltage at node N₁ is low, so transistor T₁ is also off. Current from source I₁′ charges the capacitor C₁ from 0 to a reference voltage V_(ref) (e.g., 1.2 V). The capacitor C₁ charges up and when it reaches V_(ref) that stage is selected by the maximum selector and the transistor T₁ turns on. Current from the source I₁ pulls up the voltage at J₇, which causes the negative feedback loop to turn on secondary reference transistor T_(ref)′ more in order to compensate the additional current. With the current flowing through T_(ref)′ higher than the current supplied by supply I₁, the voltage at the drain D₁ of the transistor T₁ is pulled down, which pulls the input of the inverter INV₁ low. As a result, the output of the inverter INV₁ is high, which sets the latch L₁ and resets the latch L₂. This turns on the switch SW₁ and turns off the switch SW₂.

Similarly, when the switch SW₂ is off (and switch SW₁ is latched on), the current source I₂′ is charging the capacitor C₂ from 0 to V_(ref) (e.g., 1.2 V). When the voltage on the capacitor C₂ reaches V_(ref), the transistor T₂ turns on. The reference transistor T_(ref)′ supplies more current than the source I₁ so that pulls drain D₂ of the transistor T₂ down. The input of the INV₂ goes low and the output of INV₂ goes high, which sets the latch L₂, turning on the switch SW₂ and resetting the latch L₁. The output of the latch L₁ therefore goes low and the switch SW₁ turns off.

As may be seen from the foregoing, when the voltage at node N₂ turns on transistor T₂, the voltage at N₁ is turns off transistor T₁ and vice versa. Consequently, the oscillator 300 or 310 can produce oscillation signals in two different phases. FIGS. 3C-3D show simulated signal profiles at nodes N₁ and N₂ in the electrical circuits shown in FIGS. 3A-3B operating at 33.3 MHz. As shown in the figures, the sawtooth waveforms are 180 degrees apart. FIG. 3E shows the two sawtooth waveforms of FIGS. 3C-3D combined together. Note that because there are two input stages, the capacitor for one stage can begin charging as soon as the capacitor for the other stage starts to discharge. Thus there is no need to wait for the capacitor to fully discharge before starting the next cycle, and the delay time 101 of FIG. 1B is eliminated. The one shot timers in this invention are optional and are only used to help provide digital signals. They do not interfere with the timing of the circuit.

Alternatively, the second reference transistor T_(ref) may be replaced by a current source such as I_(ref)′ of FIG. 2A, however this embodiment would not have negative feedback. In another alternative embodiment, as shown in the circuit 300′ of FIG. 3F, the negative feedback for the reference stage 306′ may further include an op amp OA_(ref) and a resistor R_(ref). The resistor R_(ref) is connected in series between the source of second reference transistor T_(ref)′ and ground. The output of the op amp OA_(ref) is connected to the gate of second reference transistor T_(ref)′, one input of op amp OA_(ref) is connected between second reference transistor T_(ref)′ and resistor R_(ref), and the other input of op amp OA_(ref) is connected to drain of first reference transistor T_(ref) having potential V. The supply current I_(s) flowing through second reference transistor T_(ref)′ is equal to I_(s)=V_(x)/R_(ref). As with the circuit 300 of FIG. 3A, the negative feedback detects the oscillator current need, and provides the appropriate supply current L. Otherwise the circuit 300′ is the same as the circuit 300 of FIG. 3A.

The oscillator depicted in FIGS. 3A, 3B may be extended to any number of phases simply by adding more phase stages. By way of example, and not by way of limitation, a three phase oscillator can be made by adding another phase module to the circuit of the two phase oscillator of the type depicted in FIGS. 3A-3B. FIG. 4A is an electrical diagram illustrating a three phase oscillator 400. Such an oscillator may be used, e.g., in a three phase voltage converter. The oscillator 400 includes three phase stages 402, 404, 406 and one reference stage 408. The reference stage 408 is similar to the reference stage 212 of the oscillator 210 described above in FIG. 2B, which includes two reference transistors T_(ref) and T_(ref)′. The second reference transistor T_(ref)′ is configured as described above to provide negative feedback control of the current through first reference transistor T_(ref) regardless of the current from the transistors of the phase stages.

The first and second phase stages 402 and 404 are similar to the first and second phase stages 302 and 304 of the two phase oscillator 300 as described in FIG. 3A. The first phase stage 402 may include a n-channel MOS transistor T₁ having a drain D₁, a gate G₁ and a source S₁. The drain D₁ is coupled to a current source I₁ of, e.g., about 1 uA through a junction J₁. The gate G₁ is coupled to a first node N₁. The source S₁ is coupled to junction J₅ in the second phase stage 404, a junction J₆ in the third phase stage 406 and connected to a junction J₇ in the reference stage 408. The first phase stage 402 also includes a capacitor C₁ having a first terminal coupled to secondary current source I₁′ through the first node N₁ and a second terminal is connected to the ground. A switch SW₁, which can be a MOS transistor, is coupled to the capacitor C₁ with a source grounded and having a gate coupled to an output of a first latch L₁. The first phase stage 402 may further include a first inverter INV₁ having an input coupled to the current source I₁ at the junction J₁ and an output coupled to an input of the first latch L₁. A one-shot timer OS₁ may optionally be coupled between the first inverter INV₁ and the first latch I₄.

The second phase stage 404 includes a MOS transistor T₂ having a drain D₂ coupled to a current source I₂ of, e.g., about 1 uA through a junction J₂, a gate G₂ coupled to secondary current source I₂′ through a second node N₂, and a source S₂ coupled to the junction J₅. The second phase stage 404 also includes a capacitor C₂ having a first terminal coupled to the second node N₂ and a second terminal is connected to the ground. A switch SW₂, which can be a MOS transistor, is coupled to the capacitor C₂, and includes a source connected to the ground and a gate coupled to an output of a second latch L₂. The second phase stage 404 may further include a second inverter INV₂ having an input coupled to the current source I₂ at the junction J₂ and an output coupled to an input of the second latch L₂. A one-shot timer OS₂ may optionally be coupled between the second inverter INV₂ and the second latch L₂.

The third phase stage 406 includes a MOS transistor T₃ having a drain D₃, gate G₃, and a source S₃. The drain D₃ is coupled to a current source I₃ of, e.g., about 1 uA through a junction J₃. The gate G₃ is coupled to secondary current source I₃′ through a third node N₃. The source S₃ is coupled to the junction J₆. The third phase stage 406 also includes a capacitor C₃ of equal capacitance to the capacitors C₁, C₂. The period for each phase is determined by the values of the secondary current sources I₁′, I₂′, I₃′, and of the capacitors C₁, C₂, C₃ since they determine how fast the voltage across the capacitors increase. For a symmetrical oscillator with equal phases, the values of the current sources and of the capacitors should be equal. However, if unequal phases are desired, those values may be altered accordingly. The capacitor C₃ has a first terminal coupled to the third node N₃ and a second terminal is connected to the ground. A switch SW₃, which can be a MOS transistor, is coupled to the capacitor C₃, and includes a source connected to the ground and a gate coupled to an output of a third latch L₃. The third phase stage 406 may further include a third inverter INV₃ having an input coupled to the current source I₃ at the junction J₃ and an output coupled to an input of the third latch L₃. A one-shot timer OS₃ may optionally be coupled between the third inverter INV₃ and the third latch L₃.

In general, it is not necessary for the reference current I_(ref) to be equal to I₁, I₂, and I₃. It is also not also necessary for I₁′=I₂′=I₃′. However, if a symmetrical oscillator is desired, then the secondary current sources should be equal, I₁′=I₂′=I₃′, as should be the capacitances of capacitors C₁=C₂=C₃.

As shown in FIG. 4A, the output of the one-shot timer OS₁ is connected to a set S of the latch L₁ and a reset R of the latch L₂. The output of the one-shot timer OS₂ is connected to a set S of the latch L₂ and a reset R of the latch L₃. The output of the one-shot OS₃ is connected to a set S of the latch L₃ and a reset R of the latch L_(f). The one-shot timers OS₁, OS₂, OS₃ can provide digital output signals if needed. If the one-shot timers OS₁, OS₂, OS₃ are not used, the outputs of the inverters INV₁, INV₂, INV₃ may be directly coupled to the latches.

The reference stage 408 may operate in a manner similar to that described above with respect to FIG. 2B. The reference stage 408 may operate with or without feedback control of the current. In the example shown in FIG. 4A, the second reference transistor T_(ref)′ provide negative feedback control of the current through the junction J₇ from the transistors T₁, T₂, T₃, and T_(ref).

The speed of oscillation of the oscillator 400 is regulated by the current supplied by sources I₁′, I₂′, I₃′ and size of capacitors C₁, C₂, C₃. The oscillator 400 may operate as follows. At an initial stage, the switch SW₁ is off and the switches SW₂ and SW₃ are on. The voltage at the first node N₁ is ramped up by the current from the current source I₁′ until it reaches the reference voltage V_(ref), which turns on transistor T₁ and pulls the drain D₁ of transistor T₁ down. This sets the latch L₁ and resets the latch L₂. Therefore the switch SW_(f) turns on and the switch SW₂ turns off; the switch SW₃ remains latched on.

Similarly, with switch SW₂ off, when the voltage at the node N₂ goes above V_(ref) due to the current from the source I₂′, the drain D₂ of MOS transistor T₂ is pulled down. This sets the second latch L₂ and resets the third latch L₃. As a result, the second switch SW₂ turns on and the third switch SW₃ turns off. When the voltage at the node N₃ goes above V_(ref) as a result of the current from the current source I₃′, the voltage on the drain D₃ of the transistor T₃ is pulled down. This sets the latch L₃ and resets the latch L_(f). As a result, the switch SW₃ turns on and the switch SW₁ turn off, which is back to the initial stage. The three phases of the oscillator 400 correspond to the voltages at the nodes N₁, N₂ and N₃. To the oscillation frequency the capacitors C₁, C₂ or C₃ may be decreased and the current source I₁′, I₂′ and I₃′ should be increased.

FIGS. 4B-4D show simulated signal profiles at nodes N₁, N₂ and N₃ in the electrical circuits shown in FIG. 4A. As shown in the figures, the sawtooth waveforms are 120 degrees apart. FIG. 4E shows the sawtooth waveforms of FIGS. 4B-4D superimposed on each other.

A four phase oscillator may be made by adding another phase module to the circuit of the three phase oscillator of the type depicted in FIG. 4A. FIG. 5A is an electrical diagram illustrating a three phase oscillator 500, which includes four phase stages 502, 504, 506, 508 and one reference stage 510. The reference stage 510 is similar to the reference stage 212 of the oscillator 210 described above in FIG. 2B, which includes two reference transistors T_(ref) and T_(ref)′, which may be MOS transistors. The first three phase stages 502, 504, 506 are configured substantially the same as stages 402, 404, and 406 as shown in FIG. 4A with a few minor differences. Specifically, junction J₆ of the third reference stage 506 is coupled to a similar junction J_(g) in the fourth reference stage 508, which is coupled to the junction J₇ in the reference stage. Also the output of one-shot timer OS₃ is connected to the reset R of latch L₄ and the reset R of latch L₁ is connected to the output of one-shot timer OS₄.

The fourth phase stage 508 may include transistor T₄, e.g., a MOS transistor, having a drain D₄, a gate G₄ and a source S₄. The drain D₄ is coupled to a current source I₄ of, e.g., about 1 uA through a junction J₄ The gate G₄ is coupled to secondary current source I₄′ through a fourth node N₄, and the source S₄ coupled to the junction J₈. The fourth phase stage 508 also includes a capacitor C₄, which is generally of equal capacitance to the capacitors C₁, C₂, C₃. The capacitor C₄ has a first terminal coupled to the fourth node N₄ and a second terminal is connected to the ground. A switch SW₄, which can be a MOS transistor, is coupled to the capacitor C₄, and includes a source connected to the ground and a gate coupled to an output of a fourth latch L₄. The fourth phase stage 508 may further include a fourth inverter INV₄ having an input coupled to the current source I₄ at the junction J₄ and an output coupled to an input of the latch L₄. A one-shot timer OS₄ may optionally be coupled between the fourth inverter INV₄ and the fourth latch L₄.

As shown in FIG. 5A, the output of the one-shot timer OS₂ is connected to a set S of the latch L₁ and a reset R of the latch L₂. The output of the one-shot timer OS₂ is connected to a set S of the latch L₂ and a reset R of the latch L₃. The output of the one-shot OS₃ is connected to a set S of the latch L₃ and a reset R of the latch L₄. The output of the one-shot OS₄ is connected to a set S of the latch L₄ and a reset R of the latch L₁. The speed of oscillation of the oscillator 500 is regulated by size of the current from sources I₁′, I₂′, I₃′, I₄′ and size of the capacitors C₁, C₂, C₃, and C₄.

The reference stage 510 may operate in a manner similar to that described above with respect to FIG. 2B. The reference stage 510 may operate with or without feedback control of the current. In the example shown in FIG. 5A, the second reference transistor T_(ref)′ provides negative feedback control of the current through the junction J₇ from the transistors T₁, T₂, T₃, T₄, and T_(ref).

The operation of the oscillator 500 is similar to the operation of the oscillator 400 as described above. At the initial stage, the first switch SW₁ is off and the switches SW₂, SW₃ and SW₄ are on. The voltage at the first node N₁ ramps up due to the current from the source I₁′ until it reaches V_(ref), which pulls the drain D₁ of transistor T₁ down. This sets the first latch L₁ and resets the second latch L₂. Therefore, the first switch SW₁ turns on and the second switch SW₂ turns off.

Similarly, when the voltage at the second node N₂ goes above the reference voltage V_(ref) due to the current from the source I₂′, the drain D₂ of the transistor T₂ is pulled down. This sets the second latch L₂ and resets the third latch L₃. As a result, the second switch SW₂ turns on and the third switch SW₃ turns off.

When the voltage at the third node N₃ goes above the reference voltage V_(ref) due to the current from the source I₃′, the drain D₃ of the transistor T₃ is pulled down. This sets the third latch L₃ and resets the fourth latch L₄. As a result, the third switch SW₃ turns on and the fourth switch SW₄ turn off.

When the voltage at the fourth node N₄ goes above the reference V_(ref) as a result of the current from the source I₄′, the drain D₄ of the transistor T₄ is pulled down. This sets the fourth latch L₄ and resets the first latch L_(f). As a result, the fourth switch SW₄ turns on and the first switch SW₁ turn off, which is back to the initial stage.

The four phases can be observed at nodes N₁, N₂, N₃ and N₄. To increase frequency of the oscillator, the capacitors C₁, C₂, C₃ or C₄ should be decreased and the current sources I₁′, I₂′, I₃′ and I₄′ should be increased. The one-shot timers OS₁, OS₂, OS₃, OS₄ can provide digital output signals if needed. If the one-shot timers OS₁, OS₂, OS₃, OS₄ are not used, the outputs of the inverters INV₁, INV₂, INV₃, INV₄ may be directly coupled to the latches.

FIGS. 5B-5E show simulated signal profiles at nodes N₁, N₂, N₃ and N₄ in the electrical circuits shown in FIG. 5A at an oscillation frequency of 50 MHz. As shown in the figures, the sawtooth waveforms are 90 degrees apart. FIG. 5F shows the sawtooth waveforms of FIGS. 5B-5D superimposed upon each other.

Oscillators according to embodiments of the present invention as described above may be used for any switching systems. It is estimated that such oscillators can oscillate from very low frequency up to about 50 MHz based on low voltage CMOS process. A faster process will allow faster oscillation frequencies. Another advantage of such oscillators is that they can change their bias current as demand changes during transients. The number of phases can easily be changed as the complexity of the system changes. In addition, the one shot timers and inverters used in the phase modules take up relatively little real estate on a chip. Spread of frequency over the voltage range is only a few percent from 2.5 to 5 volts. Furthermore, the oscillators of the type described above consume low current because a conventional comparator is not required and the reference stage can vary the current as required.

While the above is a complete description of the preferred embodiment of the present invention, it is possible to use various alternatives, modifications and equivalents. Therefore, the scope of the present invention should be determined not with reference to the above description but should, instead, be determined with reference to the appended claims, along with their full scope of equivalents. Any feature, whether preferred or not, may be combined with any other feature, whether preferred or not. In the claims that follow, the indefinite article “A”, or “An” refers to a quantity of one or more of the item following the article, except where expressly stated otherwise. The appended claims are not to be interpreted as including means-plus-function limitations, unless such a limitation is explicitly recited in a given claim using the phrase “means for.” 

1. A method for generating an oscillator signal with a multiphase oscillator having a plurality of input stages and a reference stage, each input stage producing an input stage voltage and representing a phase for the oscillator, the method comprising: comparing the input stage voltages produced by each of the input stages to a reference voltage produced by the reference stage and select an input stage having a maximum input stage voltage; changing an output of the selected input stage having the maximum input stage voltage; detecting a current need of the oscillator with a negative feedback loop coupled to the reference stage; and providing an appropriate supply current to each input stage with the negative feedback loop.
 2. The method of claim 1, further comprising: upon selection of an input stage by the maximum selector, deactivating the selected input stage and activating a next input stage.
 3. The method of claim 1 wherein each input stage includes a transistor and the reference stage includes a reference transistor, wherein the maximum selector compares the voltage on a gate of each input stage's transistor to a voltage at a gate of the reference transistor.
 4. The method of claim 3 wherein the reference stage further comprises: a fixed current source in series with a reference transistor, wherein the negative feedback loop is configured to cause the reference transistor to allow a fixed current value of the fixed current source to pass through the reference transistor.
 5. The method of claim 3, wherein providing an appropriate supply current to each phase stage the reference stage further includes applying a current control to a source of the reference transistor, wherein the current control is configured to regulate a total current from the transistors in the input stages and the reference transistor in a negative feedback mode.
 6. The method of claim 1 wherein the reference stage includes a reference current source and a reference transistor having a gate coupled to a voltage reference and a drain coupled to the reference current source.
 7. The method of claim 1 wherein each of the input stages includes: a transistor having a drain coupled to a first current source, a gate coupled to a node and a source coupled to a source of the first reference transistor, a second current source coupled to the node, a capacitor having a first terminal coupled to the node and a second terminal, a switch coupled between the first and second terminals of the capacitor, and a logic block configured to latch on the switch and turn off the switch for a subsequent phase stage when a voltage at the node exceeds a reference voltage produced by the voltage reference in the reference stage.
 8. The method of claim 1 wherein the two or more input stages include just first and last stages, whereby the oscillator signal is a two phase oscillator signal.
 9. The method of claim 1 wherein the two or more input stages include just the first and last stages and an intermediate stage subsequent to the first stage and previous to the last stage, whereby the oscillator signal is a three phase oscillator signal.
 10. The method of claim 1 wherein the two or more stages include just the first and last stages, a first intermediate stage subsequent to the first stage, and a second intermediate stage subsequent to the first intermediate stage and previous to the last stage, whereby the oscillator signal is a four phase oscillator signal. 